SMPSs typically convert alternating current voltage (for example from a mains supply) into stabilised direct current (DC) voltages required for circuits inside electrical and electronic devices such as televisions and the like. A resonant power supply is an SMPS topology that has efficiency and cost advantages for power levels of 100 Watts and above. There is an increasing interest in the use of resonant power supplies for example due to the increasing success of flat panel displays such as LCD panels and plasma panels.
In a resonant power supply the output power is controlled by modulating the switching frequency of the converter. The switching frequency determines the impedance of the resonant circuit (consisting of one or more inductors and capacitors) that is present between the input and output of the converter.
The high efficiencies achievable by resonant converters are, in part, the result of the zero-voltage switching (ZVS) of the switches, typically MOSFETs, in the converter. Zero-voltage switching refers to the fact that the MOSFETs are switched on only when there is no voltage across them. Zero-voltage switching is also referred to as soft-switching. In the case of very dynamically varying power demands or in error conditions, the zero-voltage switching condition can be lost: that is the MOSFETs may be switched on while there is a voltage across them.
Two levels of severity can be distinguished: firstly, hard switching may occur without the body diode of the complimentary MOSFET conducting. In this less severe situation the MOSFET switches will be hard-switched, which decreases the efficiency of the MOSFET, and the resultant increased losses in the converter can also produce overheating after some time. Secondly, hard-switching whilst the body diode of the complimentary MOSFET is still conducting, may occur. This type of hard switching is referred to as reverse-recovery switching, and is a more severe possible consequence of losing the zero-voltage switching condition: MOSFETs have a very limited robustness for this condition, and can be damaged very quickly.
It is useful to provide protection for the resonant converter against this harmful reverse-recovery switching. This protection is also called capacitive mode protection. Such protection increases the reliability of the power supply. The chance of failure due to unforeseen conditions is significantly reduced. It also simplifies the evaluation and design of the power supply because it will be much more robust against errors and stress tests.
Conventional methods of providing such protection will now be described, in the context of a typical resonant power supply.
A resonant power supply is shown in FIG. 1. The half bridge controller 1 drives two MOSFET-switches Mhs and Mls 2 and 3 in a half-bridge configuration. That is, the switches are closed alternatively. After a switch is opened, and before the other switch is closed, there is a short time during which both switches are open: this is referred to as the non-overlap time. The half bridge controller uses the information of the output voltage Voutput to determine the switching frequency. The switching voltage from the half bridge (VHB) is fed to the resonant tank 4. A typical resonant tank comprises two inductors and one capacitor and is thus referred to as a LLC resonant tank. The properties of this resonant tank together with the amplitude and frequency of the half bridge node determine the power delivered to the output.
The operational range of such a LLC resonant power supply is depicted in FIG. 2. At a given input voltage (Vinput), the output power (Poutput) varies with the frequency of operation (fHB). With increasing frequency, the output power reaches a peak and then falls to zero. The variation of output power with frequency is shown for two input voltage conditions—that for a low input voltage is shown as curve 21, and that for a high input voltage is shown as curve 22. For an intermediate input voltage, the operational curve will lie between these two limits, and will follow a similar profile, broadly corresponding in to an inverted parabola. The maximum power point—that is, the peak in the output power—for the low input voltage curves is shown at 23, and that for the high input voltage curve is shown at 24. As the figure shows, the frequency corresponding to the maximum power point is a function of the input voltage.
The resonant tank loads the half bridge. Since the resonant tank contains inductors and a capacitor, the total impedance can be either inductive or capacitive. For high frequencies—that is to the right of the maximum power point in the figure—the inductors dominate and the total impedance is inductive. This region of operation is called the inductive mode region. Operation in the inductive mode is preferred because an inductive load on the half bridge enables efficient zero-voltage switching. For low frequencies—that is, to the left of the maximum power point in the figure—the capacitor dominates and the total impedance is capacitive. This region of operation is called the capacitive mode region. With a capacitive impedance there is no zero-voltage switching. Reverse-recovery switching can occur and in this condition, the MOSFETs damage easily. Therefore the capacitive mode of operation is not the preferred mode. The boundary between capacitive and inductive impedance is near the top of the curve, which corresponds to the resonant frequency of the loaded LLC tank. It is preferred to operate close to—but always to the right of—the top of the curve, corresponding to the maximum power point. In general, the efficiency will be close to a maximum there (although the relationship between efficiency and output power is complex).
The relationship between the voltage at the half bridge point and a switching of the MOSFETs will now be described, for each of capacitive and inductive impedance situations, with reference to FIG. 3. FIG. 3 shows typical switching sequences 31 and 32 for the high side MOSFET (2) and low side MOSFET (3) of FIG. 1 respectively. A small non-overlap time 33 (also called dead time) is introduced between the on-time of the high-side MOSFET and that of the low-side MOSFET 3. A similar dead time 33′ exists between the on-period of the low-side MOSFET 3 and that of the high-side MOSFET 2. The voltage for the half bridge point is shown, as trace 34, for the case of capacitive impedance and trace 35 for the case of inductive impedance. The current in the resonant tank (Iresonant) is shown by trace 36 and trace 37 for the capacitive and inductive impedances respectively.
When operating in inductive mode, once the high-side MOSFET 2 is switched off, the primary resonant current (Iresonant) discharges the capacitance of the half bridge from the input voltage to ground voltage, as shown in trace 35. After this discharge, the body diode of the low-side MOSFET 3 starts conducting. The voltage across the low-side MOSFET 3 is now zero and this MOSFET can be switched on without switching loss. Similarly, after the low-side MOSFET 3 is switched off, the primary resonant current charges the half bridge point, enabling the high-side MOSFET 2 to be switched on without switching loss.
In capacitive mode (where the switching frequency is below the resonant frequency) the current shown by trace 36 has the wrong polarity for zero-voltage switching. Instead of a voltage transition at the half bridge point, the body diode of the switched-off high-side MOSFET continues to conduct a current. This body diode in the high-side MOSFET is still conducting when the low-side MOSFET is switched on after the non-overlap time. Switching on of the low-side MOSFET results in a fast voltage step at the half bridge point (that is, there is no zero-voltage switching). The body diode in the high-side switch is now forced to block very quickly. However, because of reverse-recovery properties of the MOSFET, the body diode will not block immediately and a reverse current will flow. This high reverse current can trigger the parasitic bipolar transistor in the switched off high-side MOSFET to cause a short of the input voltage and a failure of one or other or both of the MOSFETs.
There is thus a need to prevent or limit reverse-recovery switching in resonant energy converters. Conventional methods of providing such protection include monitoring a signal and increasing the operation frequency of the device if required. For example the amplitude of the resonant current may be monitored, and if the amplitude exceeds a defined level the frequency of operation is increased either stepwise or by a gradual increase. This principle is in common use in resonant power supplies. This type of over-current/power protection limits further frequency reduction corresponding to moving further to the left, up the frequency/power curve and above a defined current/power value. This solution prevents capacitive mode operation in many cases, but not always. During high load steps or short circuit at the output, capacitive mode operation with reverse-recovery switching can still occur.
A further method of limiting capacitive mode operation is to monitor the current polarity of the resonant tank current; if the wrong polarity is evident at the switch-on moment, the frequency of operation is adjusted. This protection method is used in controllers, for example for lighting ballasts (such as device UBA2021 supplied by NXP Semiconductors). If the current has the wrong polarity the frequency will be increased. This method will prevent the converter from moving into the capacitive mode operation, but works only for relatively slow changing conditions. For fast changes, capacitive mode operation will still occur for some time. Another disadvantage is that the harmful reverse-recovery switching in capacitive mode is not entirely prevented; only the duration during which it will occur is limited. Since MOSFETs can fail quickly by reverse-recovery switching, this is not a fail-safe solution.
A further conventional protection method is to monitor the slope of the voltage at the half bridge node; if the voltage does not start to slope shortly after a switch off of the relevant MOSFET the operation frequency is increased. This protection method is also used in controllers for lighting ballasts (such as NXP Semiconductors' UBA2014). In the preferred inductive mode, the voltage slope starts directly after a MOSFET is switched off. If the slope does not start within a predefined (short) time the other MOSFET is forced to switch on and the switching frequency is increased to a high value. This protection method ends capacitive mode operation quickly by stepping back into an inductive mode. One disadvantage is that a harmful reverse-recovery switching cycle will still occur at least once. A second disadvantage is the (long) required time to return to normal operation at nominal switching frequency, after the frequency has been increased.
A further method of limiting capacitive mode operation is disclosed in patent application publication WO 01/78468. In this method the voltage step at the half bridge point which occurs when a MOSFET is switched on is monitored; the operational frequency is increased by an amount which depends on the amplitude of the hard switching. In the preferred inductive mode the zero-voltage switching condition will exist, and no voltage step at all will occur. However in capacitive mode or near capacitive mode, the zero-voltage switching condition is lost and hard-switching, with a voltage step, will occur. The amplitude of this voltage step is measured and the value of the voltage step determines the increase of frequency. This method will prevent the converter from moving into the capacitive mode frequency range, but is only effective for relatively slow changing conditions. For fast changes capacitive mode operation can still occur for a short period. Thus capacitive mode switching is not entirely prevented: only its duration is limited.
A further method which is used to prevent or limit capacitive mode switching is to monitor the voltage across the switch which is about to be switched on. If the voltage across the switch is higher than a predetermined fixed value, the switch-on moment of the MOSFET is delayed and the frequency is increased. For the low side switch the voltage across the switch corresponds to the half bridge voltage; for the high-side switch the voltage is the difference between the input voltage and the half bridge voltage. In zero-voltage switching conditions (that is in inductive mode) the voltage across the MOSFET is low at the moment the MOSFET is switched on. A circuit can be added to measure the voltage across the MOSFET, which thus prevents the switching on of the MOSFET as long as the voltage across it is not below a predetermined value. This protection solution is always effective at preventing reverse-recovery switching in capacitive mode and thereby protects the MOSFETs effectively against this failure mechanism. However, because the detection is related to an absolute fixed voltage level, it will only operate for a limited input voltage range. Further, the protection operates independently from the half bridge controller.